Searching for signals in a communications system

ABSTRACT

A method and system enables matched filters of a CDMA system to be simplified using a two stage search. A course stage and a fine stage jointly produce the location(s) of received signal path-rays. In a first stage, an oversampled digital signal is decimated, and the decimated signal is applied to a matched filter to eventually produce an approximate location. In a second stage, the oversampled signal is shifted based on the determined approximate location and then correlated to a generated code, and a more-exact location is selected from the outputs of the correlations. Alternatively, a shifted version of the generated code is correlated to the oversampled signal, and the more-exact location is selected from the outputs of those correlations.

BACKGROUND OF THE INVENTION

1. Technical Field of the Invention

The present invention relates in general to the field of communications,and in particular, by way of example but not limitation, to tuning tosignal path-rays in a wireless communications system such as a CodeDivision Multiple Access (CDMA) system.

2. Description of Related Art

Mobile wireless communication is becoming increasingly important forproviding safety, convenience, improved productivity, and simpleconversational pleasure to subscribers of wireless communicationsservices. One prominent mobile wireless communication option is cellularcommunication. Cellular phones, for instance, can be found in cars,briefcases, purses, and even pockets. With the proliferation of cellularphone users and the types of services offered, new wireless systemstandards are being developed to meet these demands.

For example, CDMA, Wideband-CDMA (W-CDMA), etc. are being implemented toimprove spectral efficiency and introduce new features. In CDMA orW-CDMA (jointly referred to as “CDMA” hereafter), signal fading iscombated by combining multiple received diverse signal path-rays in aRAKE receiver. Locations (in time) of the signal path-rays are firstfound by using a searcher. Subsequently, these path-rays are combined byusing a maximum ratio combiner (MRC). Searchers are conventionallyimplemented as one or more matched filters and a peak detector. Thesignal path-rays are matched to a certain pilot sequence, which resultsin peaks that indicate the locations of the various path-rays. The peakdetector detects these resulting peaks.

Realizing a searcher is a computationally complex endeavor; therefore,it is desirable to detect the path-rays only once. After detection, thepath-rays are consequently tracked as long as possible by using apath-ray tracker. The tracking is continued until the quality of thereceived signal reaches (e.g., falls) to a predetermined threshold.Thereafter, the tracking is ceased and a new search is initiated. Thecomputational complexity of a searcher results from, at least in part,the number of delay candidates that the searcher must consider in orderto locate the path-rays. The greater the number of delay candidates, thegreater the cost in terms of hardware, processing time, powerconsumption, silicon real estate, etc. Hence, there is a need for ameans to reduce the total number of delay candidates that must beconsidered by the searcher when locating the diverse signal path-rays.

SUMMARY OF THE INVENTION

The needs of the prior art are met by the method and system of thepresent invention. For example, as heretofore unrecognized, it would bebeneficial to reduce the total number of delay candidates that must beconsidered by a searcher of a receiver when locating diverse signalpath-rays. In fact, it would be beneficial if a searcher divided thematching process into coarse signal matching and fine signal matching toreduce the number of delay elements involved in computing the locationof signal path-rays.

The present invention is related, in one embodiment, to searching forsignal path-rays in a CDMA system. The invention is directed toconducting a primary coarse search for the signal path-rays to determinetheir general location(s) and thereafter to performing a secondary finesearch to determine their precise location(s).

The method and system of the present invention is directed, in general,to simplifying the matched filters in a CDMA receiver. The matchedfilters are simplified by reducing the number of delay candidates thatmust be addressed when searching for location(s) of path-rays of asignal to be received. The simplification of the matched filters isaccomplished by implementing a two-stage signal path-ray locationsearcher. A first coarse stage locates an approximate location of asignal path-ray. A second finer stage locates the signal path-ray moreprecisely. The more-exact location(s) may subsequently be forwarded to aset of rake fingers in a spread spectrum receiver.

In one embodiment, an analog received signal is oversampled in ananalog-to-digital conversion. In other words, the analog signal issampled more than once per chip. This oversampled signal is thendecimated to reduce the number of entries in the digital signal. Thedecimated signal is applied to matched filters, which may be composed ofat least one finite impulse response (FIR) filter. A peak detectordetects an approximate location from the output of the FIR filter.

The oversampled signal is shifted responsive to the determinedapproximate location(s). A code generator generates a code correspondingto expected data to be received. The shifted oversampled signal iscorrelated to the generated code, and a comparator selects themore-exact location from the results of the set of correlations. Inanother embodiment, the generated code is shifted and then correlated tothe oversampled signal. Again, a comparator selects the more-exactlocation from the results of the set of correlations.

The technical advantages of the present invention include, but are notlimited to the following. It should be understood that particularembodiments may not involve any, much less all, of the followingexemplary technical advantages.

An important technical advantage of the present invention is that itreduces the complexity of a searcher in a CDMA receiver by reducing thenumber of delay elements that the searcher must use. This consequentlyreduces power consumption and decreases the amount of silicon spaceoccupied by the searcher.

Another important technical advantage of the present invention is thatit enables searching to be effectuated using a two-stage scheme, therebysimplifying the complexity of computations associated with the firststage.

Yet another important technical advantage of the present invention isthe ability to first detect path-rays with a coarse time resolution andsubsequently determine the locations of the path-rays by tuning to themwith a better resolution.

The above-described and other features of the present invention areexplained in detail hereinafter with reference to the illustrativeexamples shown in the accompanying drawings. Those skilled in the artwill appreciate that the described embodiments are provided for purposesof illustration and understanding and that numerous equivalentembodiments are contemplated herein.

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete understanding of the method and system of the presentinvention may be had by reference to the following detailed descriptionwhen taken in conjunction with the accompanying drawings wherein:

FIG. 1 illustrates an exemplary section of a wireless communicationssystem in accordance with the present invention;

FIG. 2 illustrates exemplary transmission/reception apparatus for thewireless communications system of FIG. 1 in accordance with the presentinvention;

FIG. 3 illustrates an exemplary air interface format for an embodimentin accordance with the present invention;

FIG. 4A illustrates signal path-ray detection for an exemplaryembodiment in accordance with the present invention;

FIG. 4B illustrates signal path-ray detection for another exemplaryembodiment in accordance with the present invention;

FIG. 5A illustrates an exemplary higher-level diagram of signal path-raydetection for the exemplary embodiments of FIGS. 4A and 4B in accordancewith the present invention;

FIG. 5B illustrates another exemplary higher-level diagram of signalpath-ray detection for the exemplary embodiments of FIGS. 4A and 4B inaccordance with the present invention; and

FIG. 6 illustrates an exemplary method in flowchart form for detectingsignal path-rays in two stages in accordance with the present invention.

DETAILED DESCRIPTION OF THE DRAWINGS

In the following description, for purposes of explanation and notlimitation, specific details are set forth, such as particular circuits,logic modules implemented in, for example, software, hardware, firmware,etc.), techniques, etc. in order to provide a thorough understanding ofthe invention. However, it will be apparent to one of ordinary skill inthe art that the present invention may be practiced in other embodimentsthat depart from these specific details. In other instances, detaileddescriptions of well-known methods, devices, logical code (hardware,software, firmware, etc.), etc. are omitted so as not to obscure thedescription of the present invention with unnecessary detail.

Preferred embodiments of the present invention and its advantages arebest understood by referring to FIGS. 1-6 of the drawings, like numeralsbeing used for like and corresponding parts of the various drawings. Itshould be understood that the FIGURES reflect both the real (I) and thecomplex (Q) portions of the overall signal value(s) (I+jQ).

Aspects of the air interface for the International MobileTelecommunications 2000 (IMT-2000), a so-called third generationstandard, is used to describe an embodiment of the present invention.However, it should be understood that the principles of the presentinvention are applicable to other wireless (or wireline) communicationstandards (or systems), especially those that employ spread spectrumtechnology, such as those based on some type of Code Division MultipleAccess (CDMA) scheme, such as Direct Sequence (DS) CDMA (e.g., W-CDMA,IS-95-A, etc.), Frequency Hopped (FH) CDMA, time-dodging CDMA,Frequency-Time Dodging (F-TD) CDMA, etc., all of which are generallyreferred to herein as CDMA.

Referring now to FIG. 1, an exemplary section of a wirelesscommunications system in accordance with the present invention isillustrated generally at 100. The (section of) wireless communicationssystem 100 includes a base station transmit/receive antenna 105, a basestation transmitter/receiver (i.e., a transceiver (TRX)) section 110,and multiple mobile stations 115 and 125. Although only two mobilestations 115 and 125 are shown in FIG. 1, it should be understood thatthe wireless communications system 100 may include more than two mobilestations. Also illustrated are transmission 120 (from the mobile station115) and transmission 130 (from the mobile station 125). As is known inthe art, reflections, delays, etc. cause multiple signals (e.g.,transmission signals 130A, 130B, and 130C) of a transmission (e.g., thetransmission 130) to be received by the base station transmit/receiveantenna 105 and subsequently processed by the base station TRX section110.

Referring now to FIG. 2, exemplary transmission/reception apparatus forthe wireless communications system of FIG. 1 in accordance with thepresent invention is illustrated generally at 200. Aninformation-carrying signal 205 is input to a spreader 210, whichspreads the signal 205 over a wide frequency range. The spread signal ismodulated at a modulator 215 and subsequently transmitted from anantenna 220. The antenna 220 may be, for example, an antenna of one ofthe mobile stations 115 and 125. The transmission 225 is received (inseveral different signals (e.g., signal path-rays) that arrive atvarying times) at an antenna 230. The antenna 230 may be, for example,the base station transmit/receive antenna 105. It should be noted,however, that the (receiving) antenna 230 may correspond to a mobilestation and that the (transmitting) antenna 220 may correspond to a basestation in accordance with the principles of the present invention.Thus, the two-stage searching principles of the present invention mayalso be implemented in conjunction with a receiver of a mobile station,for example.

Continuing now with reference to FIG. 2, the antenna 230 receives thetransmission 225, which may include multiple signals. The transmission225 is processed by the radio frequency (RF) part 235, which forwards asignal 240 to a rake receiver 245. The rake receiver 245 combines themultiple signals to achieve an improved signal 280, the improved signal280 is thereafter forwarded to post processing 290. The rake receiver245 includes rake fingers 255 and a combiner 275. As part of, or perhapsonly associated with, the rake receiver 245 are a searcher 250 and apath tracker 260. The searcher 250, the rake fingers 255, and the pathtracker 260 receive as input(s) the signal(s) 240, which include themultiple signals of transmission 225.

The searcher 250, in accordance with the present invention, implements atwo stage searching scheme to locate one or more of the multiple signalsof the signal(s) 240 as is described in greater detail below withreference to FIGS. 4-6. The searcher 250 communicates the determinedlocation(s) to the rake fingers 255 along line 265. Once the searcher250 has located the signal path-rays, the path tracker 260 attempts totrack them as long as is possible. The various adjustments that are madeto continue tracking the signal path-rays are communicated to the rakefingers 255 from the path tracker 260 along line 270. Using the locationand tracking information from lines 265 and 270, respectively, the rakefingers 255 extract the signal path-rays by despreading the receivedinformation in manner(s) known in the art. The extracted signalpath-rays are output to a combiner 275, which may utilize maximum ratiocombining (MRC) to produce the improved signal 280.

Referring now to FIG. 3, an exemplary air interface format for anembodiment in accordance with the present invention is illustratedgenerally at 300. In CDMA systems data is segmented into portions withpredetermined durations as specified by the given CDMA standard. Theseportions in turn are segmented into smaller and smaller parts until thesmallest part, the chip, is reached. Specifically, the segmentation ofinformation in the Broad Cast CHannel (BCCH) according to the IMT-2000standard, which is a W-CDMA standard, is illustrated at 300. A superframe 305 has a duration of 720 ms and is divided into seventy-two (72)frames 310. Each frame 310 is segmented into fifteen (15) slots 315,while each slot 315 is further segmented into ten (10) symbols 320.Ultimately, the symbols 320 are each composed of two hundred andfifty-six (256) chips 325.

Radio waves propagate a calculable distance during, each chip dependingon the duration of the chip. For example, radio waves propagateapproximately 78.0 m in a duration corresponding to one chip 325 underthe W-CDMA IMT2000 standard. In the W-CDMA IMT2000 standard, one chip325 duration is defined to be 0.26 μs long. Thus, 3·10⁸ ml/s×0.26·10⁻⁶s=78.0 m, where the quantity 3·10⁸ m/s equates to the speed of the radiowaves. Within such a distance as 78.0 m, several path-rays may arrive ata CDMA receiver. Consequently, the received data is typically digitizedby oversampling the chips in order to increase the resolution for thedetection of the arrival times of the path-rays. Although theoversampling enhances the performance of the searcher, it alsounfortunately increases the complexity thereof because the matchedfilters must consequently address a greater number of delay elements asa result of the oversampling. This complexity is disadvantageous to theextent it increases hardware requirements and/or processing time.

Referring now to FIG. 4A, signal path-ray detection for an exemplaryembodiment in accordance with the present invention is illustratedgenerally at 250A. The searcher 250A detects signal path-rays when theundecimated received data is shifted and correlated to the generatedcode. Referring now to FIG. 4B, signal path-ray detection for anotherexemplary embodiment in accordance with the present invention isillustrated generally at 250B. The searcher 250B detects signalpath-rays when the generated code is shifted and correlated to theundecimated received data.

As noted above, in order to increase the resolution for the detection ofthe path-rays' arrival times in a CDMA system, the received data ispreferably oversampled several (e.g., at least more than one) times perchip. The oversampling rate may be defined as the number of times perchip that a received signal is sampled. This oversampling causes a needfor increased complexity of the matched filter(s) because more delayelements are required for the implementation. In accordance with theprinciples of the present invention, however, this increased complexityis circumvented (e.g., reduced) by dividing the matching process/deviceinto two (2) stages: coarse signal matching (denoted “Stage 1”) and finesignal matching (denoted “Stage 2”).

Continuing now with the searchers 250A and 250B of FIGS. 4A and 4B,respectively, the coarse signal matching (“Stage 1”) is described. Onepurpose of the coarse signal matching is to locate the signal path-raysapproximately. First, however, the incoming signal transmission(s) 225(of FIG. 2) are converted from analog-to-digital using an A/D converter405 by oversampling several times per chip. This A/D converter 405 may,for example, be part of the RF part 235, the rake receiver 245, or someother part (not shown). (Therefore, signal 240 may be always digital(e.g., if the A/D converter 405 is part of the RF part 235 (of FIG. 2))or may be analog at one point and digital at a later point (e.g., if theA/D converter 405 is part of the rake receiver 245).) The (now digital)signal 240 is decimated at a decimation part 410 in order to produce adecimated signal 415, which has fewer elements as compared to the numberof elements of which the signal 240 is composed. The decimated signal415 is then applied to the matched filters 420. The matched filters 420may be matched to the pilot signal of the signal transmissions) 225.

The matched filters 420 may employ at least one FIR filter 425 to locatethe signal path-rays approximately. Alternatively, it could, forexample, employ a bank of correlators, etc. The decimated signal 415(e.g., instead of the (digital) signal 240) is provided to the FIRfilter 425 to reduce the total number of delay elements to be addressedby the FIR filter 425. The approximate location of the signal path-raysmay be determined by applying a peak detector 427 to the output of thematched filters 420. The matched filters 420 of the coarse signalmatching produce a detected approximate location 460 (e.g., from theoutput of the peak detector 427).

The decimation factor for the decimation part 410 is preferably equal toor less than the oversampling rate of the A/D converter 405. If thedecimation factor is less than the oversampling rate, then the FIRmatched filters are still able to detect the signal path-rays with aresolution that is higher than the chip resolution. If, on the otherhand, the decimation factor is equal to or larger than the oversamplingrate, the filter detects the signal path-rays with a resolution that isequal to or less than, respectively, the chip resolution.

Continuing now with the searchers 250A and 250B of FIGS. 4A and 4B,respectively, the fine signal matching (“Stage 2”) is described. Thefine signal matching is performed using the approximate location(s) ofthe signal path-rays detected by the coarse signal matching Theapproximate location(s) is (are) provided as one or more delaycandidates (as represented by “D”) from the coarse signal matching. Acode generator generates a pattern of the expected data at theseapproximate location(s), and this generated code pattern is correlatedto the undecimated received data having the (over)sampled resolution. Inone exemplary embodiment (e.g., as illustrated in the searcher 250A ofFIG. 4A), the exact location(s) of the signal path-rays are detectableby shifting the undecimated received data having the (over)sampledresolution and then correlating to the generated code pattern. The exactlocation of the signal path-rays is determinable by comparing theresulting correlation values. In another exemplary embodiment (e.g., asillustrated in the searcher 250B of FIG. 4B), the exact location(s) ofthe signal path-rays are detectable by shifting the generated codepattern and then correlating to the undecimated received data having the(over)sampled resolution. The exact location of the signal path-rays isdeterminable by comparing the resulting correlation values, the selectedone(s) of which may be forwarded as output(s).

Continuing now with FIGS. 4A and 4B, the fine signal matching (“Stage2”) is performed using the detected approximate location(s) 460 (e.g.,the delay candidate(s) “D”) of the signal path-rays received from thecoarse signal matching (“Stage 1”). A code generator 435 generates apattern of the expected data as generated code data 440. With referencenow only to FIG. 4A, the detected approximate location(s) 460 (“D”) andthe (over)sampled signal 240 are applied to shifters 430(D−M/C) . . .430(D) . . . 430(D+M/C), which delay (e.g., by shifting) the(over)sampled signal 240 from “−M/C” to “+M/C” units. The unit “C”, asexplained further hereinbelow with reference to Tables 1-3, relates tothe (sub)chip resolution. More specifically, in certain embodiment(s),“C” is proportional to the inverse of the (sub)chip resolution. Forexample, if a particular embodiment operates on quarter chip resolution,then “C” is equal to four (4) in that particular embodiment. Theshifters 430(D−M/C) . . . 430(D) . . . 430(D+M/C) produce as output theshifted (over)sampled signals 400(D−M/C) . . . 400(D . . . 400(D+M/C).The shifted (over)sampled signals 400(D−M/C) . . . 400(D) . . . 400(D+M/C) and the generated code data 440 are correlated in thecorrelation elements 445. With reference now only to FIG. 4B, thedetected approximate location(s) 460 (“D”) and the generated code data440 are applied to shifters 430(D−M/C) . . . 430(D) . . . 430 (D+M/C),which delay (e.g., by shifting) the generated code data 440 from “−M/C”to “+M/C” units. The shifters 430(D−M/C) . . . 430(D) . . . 430(D+M/C)produce as output the shifted generated code data 460(D−M/C) . . .460(D) . . . 460(D+M/C). The shifted generated code data 460(D−M/C) . .. 460(D) . . . 460(D+M/C) and the (over)sampled signal 240 arecorrelated in the correlation elements 445.

Continuing now jointly with the searchers 250A and 250B of FIGS. 4A and4B, respectively, the values to be correlated (e.g., the shifted(over)sampled signals 400(D−M/C) . . . 400(D) . . . 400(D+M/C) and thegenerated code data 440 in the searcher 250A, and the shifted generatedcode data 460(D−M/C) . . . 460(D) . . . 460(D+M/C) and the (over)sampledsignal 240 in the searcher 250B) are applied to the correlation elements445. Specifically, associated with each one of the shifters 430(D−M/C) .. . 430(D) . . . 430(D+M/C) is a mixing detector 445(D−M/C)′ . . .445(D)′ . . . 445(D+M/C)′ (e.g., which may be a multiplying mixer, etc.)that receives the values to be correlated. Correlation is accomplishedby applying the output(s) of the mixing detectors 445(D−M/C)′ . . .445(D)′ . . . 445(D+M/C)′ to a corresponding set of (i) coherentintegrators 445(D−M/C)″ . . . 445(D)″ . . . 445(D+M/C)″ (e.g., each ofwhich may be a low-pass or bandpass quenchable narrow-band filter,etc.), (ii) magnitude-taking parts 445 (D−M/C)′″ . . . 445(D)′″ . . .445 (D+M/C)′″, and (iii) non-coherent integrators 445(D−M/C)″″ . . .445(D)″″ . . . 445(D+M/C)″″.

The magnitude-taking parts 445(D−M/C)′″ . . . 445(D)′″ . . .445(D+M/C)′″ take the magnitude of the signal if n=1, the magnitudesquared if n=2, etc. The magnitude-taking part is used to enablenon-coherent integration by taking away the phase of the signal.Consequently, robust integration may be achieved because phasevariations in the channel do not affect the result. This protection fromphase variations can be accomplished, for example, by squaring thesignal (when n=2) or by merely taking the magnitude (when n=1). Thelatter (i.e., magnitude-taking) is advantageously cheaper to implementin terms of silicon area and power consumption while the former (i.e.,squaring) advantageously provides slightly better performance. Thecorrelation values 450(D−M/C) . . . 450(D) . . . 450 (D+M/C) are outputfrom the non-coherent integrators 445(D−M/C)″″ . . . 445(D)″″ . . .445(D+M/C)″″. A comparison part 455 selects the highest correlationvalue from among the correlation values 450(D−M/C) . . . 450(D) . . .450(D+M/C) and forwards it as a more-exact, fine location output on line265. The comparison part 455 may select the correlation value from amongthe correlation values 450(D−M/C) . . . 450(D) . . . 450(D+M/C) that hasthe largest value. Alternatively, a more-complicated scheme, forexample, may be employed to choose the best candidate.

Referring now to FIG. 5A, an exemplary higher-level diagram of signalpath-ray detection for the exemplary embodiments of FIGS. 4A and 4B inaccordance with the present invention is illustrated generally at 500.The searcher 500 operates in parallel. Referring now to FIG. 5B, anotherexemplary higher-level diagram of signal path-ray detection for theexemplary embodiments of FIGS. 4A and 4B in accordance with the presentinvention is illustrated generally at 550. The searcher 550 operates inseries. Each of the searchers 500 and 550 begin with “Stage 1” (asidentified above with reference to FIGS. 4A and 4B) blocks 505 and 555,respectively. Each of the searchers 500 and 550 include one or more“Stage 2”s. It should be noted that “Stage 2” for the searchers 500 and550 need not include the comparison parts 455 of the searchers 250A and250B (of FIGS. 4A and 4B, respectively) because their function may beaccomplished by the comparison parts 515 and 570 of the searchers 500and 550, respectively.

“Stage 1” blocks 505 and 555 produce a number of delay candidates D₁ . .. D_(k). The value of “k” may be, for example, five (5) or six (6). Inthe searcher 500, the delay candidates D₁ . . . D_(k) are produced bythe “Stage 1” block 505 approximately simultaneously and sent as avector to the “Stage 2” (as identified above with reference to FIGS. 4Aand 4B) blocks 510. The “Stage 2” blocks 510(1) . . . 510(k) eachproduce an output for a total of “k” outputs that are subsequentlycompared in the comparison part 515, which also receives as input thedelay candidates D₁ . . . D_(k). In the searcher 550, the delaycandidates D₁ . . . D_(k) are produced by the “Stage 1” block 555approximately simultaneously and sent as a vector to the “Stage 2” block560. The “Stage 2” block 560 is operated repeatedly (e.g., in serial)“k” times. The serially-produced “k” outputs of the “Stage 2” block 560are placed in a memory 565 in locations 1 . . . k, respectively. Becauseeach of these “k” outputs actually include “2M+1” (sub)outputs, eachmemory location 1 . . . k of the memory 565 may contain “2M+1” memoryslots. These “k” outputs (or, more precisely, these “k*(2M+1)” outputs)are then passed in parallel to the comparison part 570, which alsoreceives as input the delay candidates D₁ . . . D_(k).

With respect to both searchers 500 and 550, these “k” (or “k*(2M+1)”)outputs from either the multiple “Stage 2” blocks 510(1) . . . 510(k) orthe single “Stage 2”, block 560 (e.g., via the memory 565) are comparedin comparison parts 515 and 570, respectively. The comparison parts 515and 570 may, for example, select the “L” largest of the “k” (or“k*(2M+1)”) outputs that correspond to delay candidates that are themost significant path-rays by, e.g., studying their amplitudes,especially those that are more than one-half chip apart, as is explainedhereinbelow in greater detail with reference to Table 3. These selected“L” outputs may be employed in a rake receiver (e.g., the rake receiver245 of FIG. 2) in order to combine the corresponding signal-path raysusing, for example, MRC.

An exemplary comparison for the comparison parts 515 and 570 is nowdescribed with reference to Tables 1-3 for explanatory, but notlimiting, purposes. Assume that the intention is to locate two peaks(e.g., “L=2”) using two (2) “Stage 2” blocks (e.g., two “Stage 2” blocks510(1) and 510(2) or the single “Stage 2” block 560 operated twice) witheach “Stage 2” block functioning at a quarter chip resolution (e.g.,“C=4”). Considering the case when “M=2”; (and therefore each “Stage 2”has “2M+1” outputs), the number of correlators and thus outputs perstage is equal to five (5). In the Table 1 below, the output of apreceding “Stage 1” block 505 or 555 is given as [1, 2]. Theconsequential outputs of the two “Stage 2” blocks are therefore:

TABLE 1 (L = 2; M = 2; C = 4; First Stage Output [1,2]). Stage 2:1 Stage2:2 Correlator 1 D1 − 2/C = 0.5 D2 − 2/C = 1.5 Correlator 2 D1 − 1/C =0.75 D2 − 1/C = 1.75 Correlator 3 D1 = 1.0 D2 = 2.0 Correlator 4 D1 +1/C = 1.25 D2 + 1/C = 2.25 Correlator 5 D1 + 2/C = 1.5 D2 + 2/C = 2.5

In another example, consider that the output of the “j”^(th) correlatorof the “i”^(th) “Stage 2” block is denoted as OUT(i,j) as in Table 2below:

TABLE 2 (OUT (“i”^(th) “Stage 2” block, “j”^(th) correlator)). OUT (1,j)OUT (2,j) Correlator 1 140 120 Correlator 2 121 80 Correlator 3 70 30Correlator 4 60 20 Correlator 5 120 10

The final delay values to be utilized by the rake receiver may beselected by observing and analyzing these exemplary values. In thisexample, assume that the objective is to select the two (2) (e.g., L=2)best delay candidates. There are many possible approaches to selectingthese two (2) best delay candidates. A straightforward approach is tofirst determine the delay value having the largest output, which is theOUT(1,1) delay candidate having a delay of 0.50 chip. Thereafter, allthe outputs closer to half a chip are set equal to zero. The Table 3below reflects this setting to zero:

TABLE 3 (OUT (“i”^(th) “Stage 2” block, “j”^(th) correlator)). OUT (1,1)= 140 OUT (2,1) = 120 OUT (1,2) = 0 OUT (2,2) = 80 OUT (1,3) = 0 OUT(2,3) = 30 OUT (1,4) = 60 OUT (2,4) = 20 OUT (1,5) = 120 OUT (2,5) = 10

From the values in Table 3, the next largest output value is selected,which is the OUT(1,5) and OUT(2,1) delay candidates, where the delay isequal to 1.5 chips. This process may be repeated if more delaycandidates are to be determined. In this example, the two “Stage 2”stages overlap at delays of 1.5 chips. It should be noted that thisoverlap may possibly be avoided by carefully adjusting the delays whenthey are provided to the “Stage 2” stages.

It should be understood that the elements of FIGS. 2 and 4-5 need not bediscrete physical devices. They may alternatively be, for example, logicmodules in which the various functions are performed by separateentities, overlapping entities, some combination thereof, etc.Furthermore, they may also be composed of one or more software programsor routines running on a general purpose microprocessor, such as adigital signal processor (DSP), or a specialized processing unit. Otherpossibilities for realizing the principles of the present invention willbecome apparent to those of ordinary skill in the art after reading andunderstanding this disclosure, especially with regard to FIGS. 2 and 4-6and the text related thereto.

Referring now to FIG. 6, an exemplary method in flowchart form fordetecting signal path-rays in two stages in accordance with the presentinvention is illustrated generally at 600. The flowchart 600 begins withthe reception of a signal (block 605). The signal may be converted fromanalog-to-digital, preferably by oversampling multiple times per chip.The coarse signal matching stage (block 610) follows. As part of thecoarse signal matching stage (block 610), the (over)sampled signal isdecimated (block 615). The decimated signal may then be applied to afilter (block 620). The filter may be, for example, a FIR filter that ispart of a set of matched filters. The filtered signal may subsequentlybe applied to a detector to determine approximate location(s) of thesignal path-rays (block 625). The detector may, for example, be a peakdetector. It should be understood that although the present invention isdirected to dividing the searching process/device into two stages, atleast portions of the coarse and fine signal matching stages may occurin parallel.

The fine signal matching stage (block 630) may utilize the approximatelocation(s) as a guideline for shifting at least one of the values to becorrelated. In one exemplary embodiment, the undecimated (over)sampledsignal may be shifted (block 635), and the shifted undecimated(over)sampled signal may be correlated to generated code (block 645).The correlation results may then be compared, and the highestcorrelation value may be selected in order to determine the finelocation(s) of the signal path-rays (block 655). In another exemplaryembodiment, the generated code may be shifted (block 640), and theshifted generated code may be correlated to the undecimated(over)sampled signal (block 650). The correlation results may then becompared, and the highest correlation value may be selected in order todetermine the fine location(s) of the signal path-rays (block 660).After determining the fine location(s) of the signal path-rays as partof the fine signal matching stage (block 630), the fine location(s) ofthe signal path-rays may be provided to rake fingers (block 665) tofurther process the received signal.

Although preferred embodiment(s) of the method and system of the presentinvention have been illustrated in the accompanying Drawings anddescribed in the foregoing Detailed Description, it will be understoodthat the present invention is not limited to the embodiment (s)disclosed, but is capable of numerous rearrangements, modifications, andsubstitutions without departing from the spirit and scope of the presentinvention as set forth and defined by the following claims.

1. A method for locating signal path-rays in a communications system,comprising the steps of: receiving a signal; decimating said signal toproduce a decimated signal; processing said decimated signal to produceat least one first location having a first precision; processing saidreceived signal and a generated code using said at least one firstlocation having said first precision to produce at least one secondlocation having a second precision, said first precision being lessprecise than said second precision; and said step of processing saidsignal and a generated code using said at least one first location toproduce at least one second location comprises the step of shifting oneof said signal and said generated code responsive to said at least onefirst location to create plurality of shifted variables.
 2. The methodaccording to claim 1, wherein said step of processing said signal and agenerated code using said at least one first location to produce atleast one second location further comprises the step of correlating saidplurality of shifted variables with said non-shifted one of said signaland said generated code to produce a plurality of correlation values. 3.The method according to claim 2, wherein said step of processing saidsignal and a generated code using said at least one first location toproduce at least one second location further comprises the step ofcomparing said plurality of correlation values to select said at leastone second location.
 4. The method according to claim 2, wherein saidshifted variables comprise said signal and said non-shifted one of saidsignal and said generated code comprises said generated code.
 5. Themethod according to claim 2, wherein said shifted variables comprisesaid generated code and said non-shifted one of said signal and saidgenerated code comprises said signal.
 6. A receiver system for locatingsignal path-rays in a communications system, comprising: a decimationpart that decimates a signal in accordance with a decimation factor; atleast one filter connected to said decimation part, said at least onefilter involved in determining a first location of said signal; a codegenerator part, said code generator part adapted to generate a codepattern, wherein a version of said code pattern is an un-shifted versionof said code pattern; at least two shifters connected to said at leastone filter to receive said first location and said signal, said at leasttwo shifters for shifting said signal to produce at least two shiftedversions of said signal based on said first location; and at least twocorrelators, of each said at least one correlator correlating the atleast two shifted versions of said signal to the un-shifted version ofsaid at least one code pattern.
 7. The receiver system according toclaim 6, further comprising an analog-to-digital converter, saidanalog-to-digital converter converting said signal to a digital sampledsignal prior to said decimation part decimating said signal.
 8. Thereceiver system according to claim 7, wherein a sampling rate of saidanalog-to-digital converter is such that an analog version of saidsignal is sampled a plurality of times per chip.
 9. The receiver systemaccording to claim 8, wherein said sampling rate and said decimationfactor are determinative, at least in part, of a precision of said firstlocation.
 10. The receiver system according to claim 6, furthercomprising a peak detector; and wherein said at least one filterincludes at least one finite impulse response (FIR) filter, an input ofsaid peak detector is comprised of an output of said at least one FIRfilter, and said first location is comprised of an output of said peakdetector.
 11. The receiver system according to claim 6, wherein each ofsaid at least two of correlators including a multiplying mixer and anintegrator.
 12. The receiver system according to claim 6, furthercomprising a comparison part; and wherein each of said at least two ofcorrelators outputs a correlation value, said comparison part selectsthe highest value from among the output correlation values, and a secondlocation output from said comparison part is comprised of said highestvalue.
 13. The receiver system according to claim 12, wherein a firstprecision of said first location is less exact than a second precisionof said second location.
 14. The receiver system according to claim 6,wherein said communications system comprises a wireless Code DivisionMultiple Access (CDMA) communications system.
 15. The receiver systemaccording to claim 6, further comprising a comparison part and aplurality of rake fingers, said comparison part receiving an output fromof each said at least two correlators and providing a second location tosaid plurality of rake fingers.
 16. A method for locating signalpath-rays in a communications system, comprising the steps of: receivinga signal; decimating said signal to produce a decimated signal;processing said decimated signal to produce at least one first locationhaving a first precision; processing said received signal and agenerated code using said at least one first location having said firstprecision to produce at least one second location having a secondprecision, said first precision being less precise than said secondprecision; said step of processing said signal further comprising thesteps of: generating said code; shifting based on said first location;correlating said generated code to said signal, one of said generatedcode and said signal having been shifted in said step of shifting; andselecting said second location in response to said step of correlating.17. A receiver system for locating signal path-rays in a communicationssystem, comprising: a decimation part that decimates a signal inaccordance with a decimation factor; at least one filter connected tosaid decimation part, said at least one filter involved in determining afirst location of said signal; a code generator part, said codegenerator part adapted to generate a code pattern; at least two shiftersconnected to said at least one filter to receive said first location,said signal or said code pattern, said at least one shifter performingeither: a shifting of said signal to produce a shifted version of saidsignal based on said first location; or a shifting of said code patternto produce a plurality of shifted versions of said code pattern based onsaid first location; at least two correlators, each of said twocorrelators performing at least one of: correlating said shifted versionof said signal to an un-shifted version of said code pattern to producea second location, when said signal is shifted by said shifters; andcorrelating an un-shifted versions of said signal to said shiftedversion of said code pattern to produce said second location when saidcode sequence is shifted by said shifters.
 18. A receiver system forlocating signal path-rays in a communications system, comprising: adecimation part that decimates a signal in accordance with a decimationfactor; at least one filter connected to said decimation part, said atleast one filter involved in determining a first location of saidsignal; a code generator part, said a code generator part adapted togenerate code pattern; at least two shifters connected to said at leastone filter to receive said first location and said code pattern, said atleast two shifters for shifting said code pattern to produce a pluralityof shifted versions of said code pattern based on said first locations;and at least two correlators, of each said at least two correlatorscorrelating an un-shifted version of said signal to the plurality ofshifted versions of said code pattern.
 19. The receiver system accordingto claim 18, further comprising analog-to-digital converter, saidanalog-to-digital converter converting said signal to a digital sampledsignal prior to said decimation part decimating said signal.
 20. Thereceiver system according to claim 19, wherein sampling rate of saidanalog-to-digital converter is such that an analog version of saidsignal is sampled a plurality of times per chip.
 21. The receiver systemaccording claim 20, wherein said sampling rate and said decimationfactor are determinative, at least in part, of a precision of said firstlocation.
 22. The receiver system according to claim 18, furthercomprising a peak detector; and wherein said at least one filterincludes at least one finite impluse response (FIR) filter, an input ofsaid peak detector is comprised of an output of said at least one FIRfilter, and said first location is comprised of an output of said peakdetector.
 23. The receiver system according to claim 18, wherein each ofsaid at least two correlators including a multiplying mixer and anintegrator.
 24. The receiver system according to claim 18, furthercomprises a comparison part; and wherein each of said at least two ofcorrelators outputs a correlation value, said comparison part selectsthe highest value from among the output correlation values, and a secondlocation output from said comparison part is comprised of said highestvalue.
 25. The receiver system according to claim 24, wherein a firstprecision of said first location is less exact than a second precisionof said second location.
 26. The receiver system according to claim 18,wherein said communications system comprises a wireless Code DivisionMultiple Access (CDMA) communications system.
 27. The receiver systemaccording to claim 18, further comprising a comparison part and aplurality of rake fingers, said comparison part receiving an output fromeach of said at least two correlators and providing a second location tosaid plurality of rake fingers.